Device and method for inductive measurements - self test

ABSTRACT

A method for nondestructive and noncontact detection of faults in a test piece, with a transmitter coil arrangement with at least one transmitter coil, a receiver coil arrangement with at least one receiver coil for detecting a periodic signal which has a carrier oscillation whose amplitude and/or phase is modulated by a fault in the test piece. A signal processing unit producing a useful signal receiver coil signal, and an evaluation unit evaluating the useful signal for detection of a fault in the test piece. A self-test unit automatically or upon an external request undertakes systematic quantitative checking of signal processing functions of the signal processing unit and/or systematic quantitative checking of the transmitter coil arrangement and/or of the receiver coil arrangement and/or upon external request undertakes calibration of the signal processing unit using a calibration standard which replaces the transmitter coil arrangement and/or of the receiver coil arrangement.

CROSS REFERENCE TO RELATED APPLICATION

This application is a divisional of co-pending U.S. patent applicationSer. No. 12/783,693.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a device and method for nondestructive andnoncontact detection of faults in a test piece. In particular, thepresent invention relates to fault detection using measurements of aneddy current or stray magnetic flux. Furthermore, the invention relatesto a device and a method for detecting electrically conductive particlesin a liquid flowing in a pipe segment using the eddy currents induced inthe particles being detected.

2. Description of Related Art

Conventional nondestructive and noncontact fault detection of faults ina test piece of a semi-finished metallic product is performed bymeasuring induction and eddy currents in the test piece. In doing so,the test piece is exposed to periodic alternating electromagnetic fieldsthrough a sinusoidally energized transmitter coil. The resulting eddycurrents induced in the test piece in a coil arrangement are used as aprobe and induce a periodic electrical signal, which has a carrieroscillation according to the transmitter carrier frequency whoseamplitude and/or phase is modulated by a fault in the test piece when afault travels into the sensitive region of the probe. Conventionally,when scanning the test piece, the test piece is moved linearly withrespect to the probe; however, arrangements with a rotating probe alsoknown. For example, an eddy current measurement device with a linearlyadvanced test piece is described in U.S. Pat. No. 5,175,498.

Similarly, electrically conductive particles in a liquid, which flowsthrough the coils, cause eddy current losses. These eddy currents can bedetermined by measuring the impedance change of the coils. In this wayelectrically conductive particles in a liquid flowing in a pipe can bedetected by means of an inductive coil arrangement. This is especiallyadvantageous for detection of the concentration of metallic particles inthe lubricant circuit of a machine in order to draw conclusions aboutthe machine state such as measurements of machine wear.

Another conventional measurement method for nondestructive andnoncontact detection of faults in a test piece is stray magnetic fluxmeasurement (or stray magnetic field measurement), by means of aninduction coil with a magnetic yoke, which magnetizes the test pieceresulting in a stray magnetic flux produced by the test piece. Themagnetic flux is measured by means of a suitable sensor. Faults in thetest piece are detected based on their effects on the stray magneticflux. One example of this stray flux measurement can be found in U.S.Pat. No. 4,445,088.

In eddy current measurement devices containing probes which rotatearound the periphery of the test piece, measuring the distance betweenthe probe head and test piece is performed in order to correct themeasurement with respect to the distance because the distance fluctuatesduring the course of one revolution. The measurement is performedbecause of decentering or asymmetry of the cross section of the testpiece occurs during one revolution. One example of this arrangement canbe found in German Patent Application No. 40 03 330 A1.

International Patent Application Publication WO 2006/007826 A1 disclosesan eddy current measurement device with a digital front end, such thatthe A/D converter stage is triggered with a n-th integral fraction ofthe frequency of the carrier oscillation, where n is selected dependingon the fault frequency, i.e., the quotient of the relative velocitybetween the test piece and probe and the effective width of the probe.

U.S. Pat. No. 4,209,744 describes an eddy current measurement devicewhich has a test means which simulates signals that are typical offaults in a test piece in order to perform fundamental checking of theelectronics. However, only a single amplitude and a primary faultfrequency can be simulated. Even if the simulated fault signal wereprovided with variations, all the electronics cannot be tested.Furthermore, this simulated fault signal cannot be attributed to acertified reference element without dismounting all the electronics andsending them to a laboratory.

International Patent Application Publication WO No. 01/22075 A2describes an eddy current measurement device within the framework ofself-calibration of the system. The intensity of the signal originatesfrom a segment of a test piece without a determining a fault.

GB Patent Application No. 2 192 064 describes an inductive test devicewhere the device is detuned to simulate a fault by a self-test means andby connecting a LED.

SUMMARY OF THE INVENTION

A primary object of this invention is to devise a device and method fornondestructive and noncontact detection, especially by means of eddycurrent measurement, or stray flux measurement, of faults in a testpiece or by detecting electrically conductive particles in a liquidflowing in a pipe segment, to ensure that measurement is as reliable aspossible.

The above object of the invention is achieved in a device as describebelow.

In the approach in accordance with the invention, it is advantageousthat because the self-test unit undertakes systematic quantitativechecking of the signal processing functions of the signal processingunit, the transmitting coil arrangement, and the receiver coilarrangement, and upon request to undertake calibration of the signalprocessing unit with a calibration standard which is to replace thetransmitter coil arrangement and/or the receiver coil arrangement. Thisallows for comprehensive checking of the functions of the front-end,especially of the filters and amplifiers as well as the probe, and thus,high reliability of the measurement results is achieved. In particular,calibration of the device is also easily enabled. This appliesespecially to calibration with respect to the adjustable preamplifier.

Altogether, increased reliability of the test results is achieved sincefaults in the individual electronic components of the device can bereliably detected. In particular, high reliability is achieved comparedto the calibration known in the prior art on a simulated sample faultsince the latter in practice generally does not emerge in simulated formand thus the meaningfulness of calibration on such a sample fault isrelatively low. Further, the individual components cannot be separatelyquantitatively checked.

Instead, in the nondestructive and noncontact detection of faults in amoving test piece, i.e., in an eddy current test device or a stray fluxmeasurement device, as described herein, can be used in the detection ofelectrically conductive particles in a liquid flowing in a pipe segmentwith a velocity, such as a particle counter.

Preferably, the self-test unit is made to switch the signal processingunit for checking the signal processing functions such that the signalfor the transmitter coil arrangement is fed directly as a periodic inputsignal into the signal processing unit, the input signal systematicallyvaried. Typically the signal processing unit has amplifiers andfrequency filters, the self-test unit being made to check by means ofvariation of frequency and amplitude of the signal for the transmittercoil arrangement whether the measured gain of the amplifiers and themeasured corner frequencies and steepnesses of the frequency filters arewithin the given specification, and a corresponding fault signal isoutput if the specification is not satisfied.

Preferably, the driver for the transmitter coil has a current sensor.The self-test unit monitors and determines the impedance of thetransmitter coil from the transmitter coil current and the transmittercoil voltage and the offset voltage of the receiver coil.Advantageously, the self-test unit is made to store the transmitter coilcurrent and the receiver coil offset voltage as a function of time inorder to enable observation of long-term changes of the transmitter coiland the receiver coil.

The device can be made with several channels, the transmitter coilarrangement and the receiver coil arrangement have several coils whichare each assigned to one certain measurement frequency.

Preferably, the calibration standard is at least one RC element, and bymeans of a calibrated measurement resistance of the RC element the A/Dconverter, or converters of the signal processing unit, can be checkedwith respect to their accuracy, and the sampling frequency of theprocessor of the signal processing unit can be checked by means of thecorner frequency of the RC element. The calibration standard can be avoltage divider which has been certified by a test laboratory. Thus, thesensitivity of the entire system can be checked with a calibratedreference element so that the entire system can be checked at least witha typical setting.

Preferably, the front-end is made digital, i.e., the receiver coilsignal is sampled by means of a triggerable A/D converter stage and thenfiltered by means of frequency filters to obtain a demodulated usefulsignal. The A/D converter stage is triggered with the n-th integralfraction of the frequency of the carrier oscillation of the signal forthe transmitter coil arrangement, n is chosen depending on the faultfrequency which arises as the quotient of the relative velocity betweenthe test piece and the receiver coil arrangement and the effective widthof the receiver coil arrangement, and the frequency filters being set asa function of the fault frequency.

Typically, the signal processing unit has an adjustable preamplifier forthe receiver coil signal, and the preamplifier can be checked by thecalibration standard made as a RC element being exposed to a fixedsinusoidal voltage whose amplitude is chosen such that in the leastsensitive setting of the preamplifier a sinusoidal signal can bedigitally converted with the desired accuracy by means of the A/Dconverter stage so that at higher gains of the preamplifier, thesinusoidal signal is overdriven. The overdriven sinusoidal signal isreconstructed with a mathematic approximation, for example, by means ofa compensation computation, in order to determine the actual signalamplitude.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an inductive measurement device with aself-test function and calibration function in accordance with theinvention;

FIG. 2 is a block diagram of an aspect of the invention which is usedfor detecting faults in a moving test piece;

FIG. 3 is a block diagram of one example of an inductive measurementdevice according an aspect of to the invention which is used fordetecting electrically conductive particles in a flowing liquid;

FIG. 4 schematically illustrates a longitudinal section through a pipethrough which a liquid is flowing and which is provided with atransmitter and receiver coil for use with the measurement device asshown in FIG. 3, and

FIG. 5 is a block diagram of the wiring of the coils from FIG. 4.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 shows a block diagram of an inductive measurement device with aself-test function and calibration function according to an aspect ofthe invention. A signal processor 60 communicates with a programmablearray logic (PAL) element 68. The PAL element 68 is designed to controlthe A/D converter and D/A converter. The PAL element 68 also supplies atransmitter coil driver 70 which is provided with a current sensor 72,and delivers the signal for the transmitter coil arrangement (not shownin FIG. 1) of the probe 11 (i.e., measurement head). The receiver coilsignal of the receiver coil arrangement (not shown in FIG. 1) of theprobe 11 is provided to a low-noise amplifier 74 which is used as apreamplifier. The gain of the low-noise amplifier 74 is controlled orvariably set by the processor 60 by way of the PAL element 68. Thesignal amplified by the amplifier 74 passes through a resonance filter78 and is supplied to the PAL element 68. The processor 60 forprocessing, or evaluation, of the signal after digitization in an A/Dconverter 80, which can be designed to handle 18 bits. In this way, fromthe receiver coil signal is produced which is then evaluated by anevaluation unit. The evaluation unit can be implemented in the form ofthe processor 60 and/or externally, for example, as a personal computer(PC) 64.

Furthermore, the system can have a distance sensor 82 with a transmittercoil and a receiver coil (not shown) in order to produce a distancesignal from the receiver coil signal of the distance sensor 82. Thedistance signal constitutes a measure of the distance between the testpiece and the probe 11. There is a driver 84 for the transmitter coil ofthe distance sensor 82 which has a current sensor 86 and which issupplied by the PAL element 68. The receiver coil signal of the distancesensor 82 is supplied to a unit 88 which performs amplification, offsetand rectification of the distance signal. The unit 88, like theamplifier 74, is controlled by the PAL element 68. The distance signalis supplied to the PAL element 68, and then to the processor 60 forevaluation by an A/D converter 90, which can be designed to handle 16bits. Further, there can be several distance sensors 82.

The elements 68, 70, 74, 76, 78, 80, and optionally 60, as well aselements 84, 86, 88, 90 are part of the signal processing unit whichproduces a signal for evaluation by the evaluation unit from thereceiver coil signals.

A self-test unit 62 is implemented in the processor 60. The self-testunit 62 performs systematic quantitative checking of the signalprocessing functions of the signal processing unit of the front end andsystematic quantitative checking of the probe 11 and of the distancesensor 82. Further, the processor 60 performs the checkingautomatically, at system startup, or at the request of the userinterface.

A switch arrangement 66 with three switches 63, 67, 69 is used formonitoring the signal processing unit. The three switches 63, 67, 69 canbe actuated by the self-test unit 62 (in doing so the switches 63 and 67are opened and the switch 69 is closed) in order to feed the signal forthe transmitter coil of the probe 11 as a periodic input signal into thesignal processing unit, i.e., into the input of the amplifier 74 bybypassing the transmitter coil directly.

In the self-test, the self-test unit 62 provides the signal thetransmitter coil which is varied with respect to frequency and amplitudein order to check whether the gain of the amplifier 74 is within therequired specifications. The gain has been measured and the measuredcorner frequencies and steepness of the frequency filter 78. Acorresponding fault signal is output to the user interface 64, 65, ifthe specification is not satisfied.

According to aspects of the invention, the device can be made withseveral channels, then are provided to transmitter coil driver 70, theprobe 11 and the self-monitoring switch arrangement 66 is present foreach channel and a multiplexer 76 connected upstream of the amplifier 74(for each transmitter coil there is then its own frequency).

A self-test switch arrangement 92 is isolated between the driver 84 andthe unit 88. The self-test switch arrangement 92 has three switches 89,91, 93 which can be actuated by the self-test unit 62 (in doing so theswitches 89 and 91 are opened and the switch 93 is closed) to induce aself-test of the unit 88, or of the A/D converter 90 by a signal outputby the transmitter coil driver 84, bypassing the transmitter coil of thedistance sensor 82 being sent directly to the input of the unit 88, andby means of the self test unit 62 the frequency and amplitude of thecoil driver signal can be systematically varied.

In addition to the output signal of the unit 88, the current signal ofthe current sensor 72 and the current signal of the current sensor 86are supplied to the multiplexer 94, which is connected upstream of theA/D converter 90. The sensor current signals are supplied, in this way,to the self-test unit 62 for evaluation. The complex impedance of therespective transmitter coil can be determined and monitored by means ofthe self-test unit 62 from the transmitter coil current and thetransmitter coil voltage detected by current sensors 72 and 86. Also, afault signal can be optionally output by way of the user interface 64and 65. As illustrated in FIG. 1, the transmitter coil voltages aremeasured at the sites labelled 1 and 3 and are supplied to the PALelement by way of the multiplexer 94 and the A/D converter 90.

Furthermore, the offset voltage of the receiver coil of the probe 11 canbe monitored by means of the self-test unit 62 (Note: only differencecoils have an offset voltage, which arises in any difference coilarrangement since two coils are never exactly identical).

The offset voltage can be eliminated from the receiver signal by meansof a high-pass filter. The difference of the voltage before and afterthe high-pass filter then yields the offset voltage.

Advantageously, the self-test unit 62 is made such that the transmittercoil current and the receiver coil offset voltage are stored as afunction of allowing for observation of long-term changes of thetransmitter coils and the receiver coils. This monitoring is especiallyimportant when the system is designed as an inductive particle counterbecause the coils cannot be easily dismounted and checked.

Furthermore, the self-test unit 62 is configured such that calibrationof the signal processing electronics is enabled by means of a certifiedcalibration standard 96 which can replace the coil 11. The calibrationstandard 96 is connected to the transmitter coil driver 70 on the inputside to the multiplexer 76 and to the amplifier 74 on the output side.When the calibration standard 96 has several reference elements, suchas, different resistances, which are switched over in the course ofcalibration, the calibration standard 96 has one terminal 98, forexample an I²C, which is connected to the processor 60 and the self-testunit 62 for undertaking the corresponding switchovers of the referenceelements.

The points labeled “2” and “4” allow for direct measurement of thevoltages upstream of the input channels of the amplifier 74 and of theunit 88. Thus, it is possible to directly measure the voltage drop withthe calibration standard 96 which was set instead of the correspondingcoil, for example.

It is preferable that the calibration standard has at least one RCelement with at least one calibrated measurement resistance for checkingthe precision of the A/D converter of the signal processing electronics.The sampling frequency of the processor 60 can also be checked with theRC element. The measurement resistance of the calibration standard 96 isa lowpass filter to suppress interference. As a reference element, thecalibration standard 96 provides for a defined voltage at the input ofthe A/D converter 80 so that unwanted fluctuations of the samplingfrequency are detected.

It is preferable that calibration be performed once a year.

The calibration standard 96 may be a separate unit independent of themeasurement device and connected to the measurement device duringcalibration. This example embodiment is advantageous because thecalibration of the calibration standard can be easily checked by acertified calibration laboratory.

Alternatively, the calibration standard 96 can be made as a part of themeasurement device such as a component which is provided on a board ofthe measurement device connected in place of the corresponding coil.This example embodiment has the advantage that the measurement devicedoes not need to be opened for preparation of calibration. However, inthis case, the calibration of the calibration standard cannot bechecked.

The calibration standard 96 is helpful especially for calibration of theadjustable preamplifier 74. When the calibration standard 96 foreconomic reasons has only a single or only a few reference resistancevalues, it is possible to proceed as follows. The RC element of thecalibration standard 96 obtains a fixed sinusoidal voltage from thetransmitter coil driver 70. The fixed sinusoidal voltage is so largethat a sinusoidal signal can be digitally converted with a desiredaccuracy by means of the A/D converter 80 in the least sensitiveposition of the amplifier 74. If the gain is increased by means of thePAL element 68, the sine is cut off at some time, and the truncated sinethen can be reconstructed again via mathematical approximation, such asa compensation calculation. As a result, the actual amplitude of thesignal can be measured. The prerequisite for this method is that theelectronics used do not have a latchup effect and the input stage of theA/D converter 80 is protected against destruction by overvoltage.

The following equation of the compensation computation for a sine may beused:

A0*n+A1*[sin (x)]+A2*[cos (x)]=[yi]

A0*[sin (x)]+A1*[sin²(x)]+A2*[sin (x)*cos (x)]=[yi*sin (x)]

A0*[cos (x)]+A1*[sin (x)*cos (x)]+A2*cos²(s)]=[yi*cos (x)]

where yi is a measured value such that y(i)=A0+A1*sin(x)+A2*cos(x) andx=2*π*f*i*dt, where f indicates the frequency. The brackets stand forsummations over the running variable 1 from zero to n. Those measuredvalues which are outside the allowable range, i.e., the “truncated”values, may not be used here. The value x represents the current angle,which need not be equidistant.

By computing the amount of A1 and A2, the original amplitudeA=SQRT(A1*A1+A2*A2) and the phase offset PHI=arctan(A2/A1) are obtained.

It goes without saying that the described signal reconstruction can beused not only in the checking of the variable amplifier 74, but also inan eddy current test, if as a result of certain circumstances thereceiver coil signals arise which overdrive the A/D converter.Ultimately, the measurement range can be expanded using software by thissignal reconstruction.

The relatively simple checking of the variable amplifier 74 describedabove allows for the storage and use of correction values for therespective gain, allowing for more economical and higher qualityamplifiers.

There are resonance filters, like the resonance filter (or a combinationof highpass and lowpass) 78, allowing for operation with a variabletransmitter frequency. The most favorable sampling frequency arising asa function of the velocity of the test piece, effective coil width, andtransmitting frequency. As already described, in a self-test usingvariation of the frequency and amplitude of the input voltage, thecorner frequencies and the edge steepness of the filters can bedetermined.

Changes of the sensor hardware, especially damage, can be ascertainedearly by the described impedance measurements of the transmitter andreceiver windings using the self-test unit 62 so that test times withdamaged sensor hardware can be avoided as much as possible. As a result,measurement becomes more reliable.

The described measurement of the receiver coil offset voltage by theself-test unit 62 enables early detection of overdriving problems, forexample, in conjunction with certain test piece materials. This allowsfor preventive reactions to problems and increases in the reliability ofthe test.

The possibility of calibration of the system by means of the self-testunit 62 and the calibration standard 96 enables simple calibration ofthe system on site, eliminating the necessity of installation anddismounting of a test adapter in the system. As a result, production andservicing of the system is more economical.

The calibration standard 96, itself, if it is made as a separate unit,can of course also be calibrated at regular intervals by a certifiedcalibration laboratory, as previously described.

FIG. 2 illustrates a block diagram of an example of an inductivemeasurement device according to an aspect of the claimed invention whichis used for detecting faults in a moving test piece and a digitaldemodulation method. Aside from the self-test function and signalreconstruction, this device is described in WO 2006/007826 A1. Here, atest piece 13 in the form of a semi-finished industrial article, forexample, a slab, which is tested when it moves linearly with a variablevelocity v past the probe 11. The velocity is detected with a velocitydetector 21 which can deliver for example a signal essentiallyproportional to the velocity v. The signal can be, for example, arectangular signal (possibly also two-track in order to be able todistinguish forward and backward) which contains one pulse, for example,per 5 mm advance of the test piece 13.

The probe 11 has a transmitter in the form of a transmitter coil 18 anda receiving coil 15. With an alternating electromagnetic field with atleast one given carrier frequency, the transmitter coil 18 is used toinduce eddy currents in the test piece 13 in the receiving coil 15 andin turn induce an AC voltage which acts as a probe signal and has acarrier oscillation with the carrier frequency of the transmitter coil18. The amplitude and the phase of the probe signal is modulated by afault 23 when the fault 23 travels into the effective width WB of thereceiving coil 15. The receiving coil 15 is preferably made as adifference coil, i.e., a coil with two windings which are wound in theopposite direction, and react only to changes of the electricalproperties due to the presence of a fault 23 of the test piece.Difference coils are suitable mainly for detection of sudden changes inthe test piece 13. An absolute coil can also be used as the receivingcoil 15 which comprises several windings wound in the same direction,and suitable especially for detection of long homogeneous changes in thetest piece 13.

The voltage for the transmitter coil 18 can be produced by a binarysignal produced by a timer unit 44 and delivered to a generator 48 asthe input frequency which produces a rectangular signal and a sinusoidalsignal which travels through the curve shaper 40 and then is amplifiedby a power amplifier 42 before being sent to the transmitter coil 18.Preferably, the signal has a sinusoidal shape and in the simplest casecontains only a single carrier frequency, but measurements with severalcarrier frequencies at the same time and/or carrier signals which differdistinctly from sinusoidal oscillations is also possible. Typically thecarrier frequency is in the range from 1 kHz to 5 MHz.

Fundamentally, the transmitter coil can also be operated with a digitalsignal based on the pulse duration modulation greatly reducing the powerloss in the driver stage.

The probe signal received by the receiving coil 15 travels through abandpass filter 19 and an adjustable preamplifier 17 before beingsupplied to an A/D converter stage 35. The bandpass filter 19 is used,on the one hand, as an (anti-)aliasing filter with respect to signaldigitization by the A/D converter stage 35, and on the other hand, tomask out high frequency and low frequency noise signals. The adjustablepreamplifier 17 is used to bring the amplitude of the analog probesignal to the amplitude optimally suitable for A/D converter stage 35.

The A/D converter stage 35 has two A/D converters 32 and 34 which areconnected in parallel and have high resolution with at least aresolution of 16 bits, preferably 22 bits. It is also preferable thatthe A/D converter stage 35 is able to carry out at least 500 A/Dconversions per second. The A/D converters 32, 34 are preferably flashconverters or SAR (successive approximation register) converters.

The version with two A/D converters is one example. It is important thatthe fault signal is orthogonally sampled, which may also be performedwith only one converter.

The A/D converter stage 35 is triggered by a trigger means 37, which hasthe aforementioned timer unit 44, a cosine generator 48, a sinegenerator 46 located parallel to the sine generator 46, and a frequencydivider 30. The signal which has been generated by the cosine generator48 and which has the frequency of the carrier frequency of the supplysignal of the transmitter coil 18 is provided to the frequency divider30. The signal of the sine generator 46 which corresponds to the signalof the cosine generator 48, but with a phase-shift of 90° thereto, isalso provided to frequency divider 30. In the frequency divider 30 thesetwo signals are divided with respect to their frequency by a wholenumber n. The corresponding frequency-reduced output signal is used totrigger the A/D converter 32 and the A/D converter 34. The selection ofthe number n for the divider 30 is undertaken by a digital signalprocessor 60 depending on the fault frequency, i.e., the quotient of thecurrent velocity of the test piece v and the effective width WB of thereceiving coil 15. Preferably, n is chosen to be inversely proportionalto the main fault frequency in order for the trigger rate of the A/Dconverter stage 35 to be at least roughly proportional to the main faultfrequency. This results in that if the effective width WB in the firstapproximation is assumed to be constant, at a higher test piece velocityv and thus a high fault frequency the analog probe signal is sampledmore frequently.

Preferably, the divider 30 is made as a so-called PAL (ProgrammableArray Logic) component in order to ensure that the trigger signalsarrive synchronously, to the output signal of the cosine generator 48and the sine generator 46 without phase jitter at the A/D converterstage 35.

Due to the corresponding phase shift of the two input signals of thedivider 30, the two A/D converters 32, 34 are also triggered with afixed phase offset of 90°. In this way the analog probe signal can beevaluated in a two-component manner, i.e., with respect to amplitude andphase. It goes without saying that the phase delay between the triggersignal of the A/D converter signal 35 and the signal of the transmittercoil 18 should be as small as possible, and especially so-called phasejitter should also be avoided, i.e., the phase relations should beconstant in time as exactly as possible.

With the illustrated trigger means 37 the analog probe signal is sampledby each A/D converter 32, 34 at least once per full wave of the carrieroscillation (in this case n is equal to 1). Depending on the currentfault frequency, i.e., the velocity of the test piece v, n can be muchlarger than 1 so that sampling only in each n-th full wave of thecarrier oscillation.

As already mentioned, what matters is that sampling is takenorthogonally. When sampling is done at 0° and 90° the complex componentsof the fault signal are obtained. At 180° and 270° the same componentsare obtained, but in the inverse to those taken at 0° and 90°. Byinverting these components an average can be formed and thus anincreased sampling rate can be used. Such sampling methods haveadvantages with respect to noise and design of the input filter.

The demodulated, digital, two-channel output signal of the A/D converterstage 35 travels through a digital bandpass filter 52 of the signalprocessor 60. The digital bandpass filter 52 is used to mask out noisesignals outside the bandwidth of the fault signal. For this purpose, thecorner frequency of the highpass filter (software filter) is preferablychosen such that it is less than one fourth of the fault frequency,while the corner frequency of the lowpass filter is preferably chosensuch that it is at least twice the fault frequency to avoid masking outthe signal portions which still contain information of the fault.

The digital bandpass 52 is clocked with the sampling rate of the A/Dconverter stage 35, i.e., the trigger rate, which is an advantage thatthe corner frequencies of the bandpass. When the fault frequencychanges, i.e., when the velocity of the test piece v changes, thetrigger rate are automatically entrained with the fault frequency sincethe corner frequencies of a digital bandpass filter are proportional tothe clock rate and the clock rate is automatically adapted to the changeof the fault frequency by way of the sampling rate which is stipulatedby the trigger unit 37.

This also applies analogously when the transmission frequency has beenchanged reducing the cost of digital filtration with respect todifferent types of filter stages.

The information necessary for determining the main fault frequency withrespect to the effective width WB can be either manually input to thesignal processor 60 made available directly by the probe 11, asdescribed, for example, in European Patent Application No. 0 734 522 B1.

It goes without saying that the measurement system reacts analogously tothe change of the fault frequency caused by the constant velocity v ofthe test piece, but the receiving coil 15 is replaced by another with adifferent effective width WB.

The useful signal is obtained after filtration by the digital bandpassfilter 52 is evaluated in a known manner by an evaluation unit 50 inorder to detect and locate faults 23 of the test piece 13. For detectionboth the amplitude information and the phase information of the faultsignal is used.

In particular, for relatively large values of n, i.e., when only arelatively small number of full waves of the carrier oscillation aresampled during the sampling pauses, the transmitter coil 15 and/or theevaluation electronics, especially the signal processor 60, can beturned off or put on stand-by in order to reduce power consumption suchcapability is important especially for portable measurement devices.

In the processor 60 the self-test unit 62 for the monitoring andcalibration functions named above in conjunction with FIG. 1 areimplemented. Thus, the self-test unit 62 controls the switch arrangement66 with three switches 63, 67, 69 in order to feed the signal for thetransmitter coil 18 of the probe 11 bypassing the transmitter coil 18and the receiver coil 15 directly as a periodic input signal into thesignal processing, i.e., into the input of the bandpass filter 19.

FIGS. 3 to 5 show one example of an inductive measurement deviceaccording to an aspect of the claimed invention used to detectelectrically conductive particles in a flowing liquid using a digitaldemodulation method. Aside from the self-test function, this device isdescribed in the German patent application not published beforehand,with application number of 10 2007 039 434.0 and corresponding to U.S.Patent Application Publicatin No. 2009/0051350. Fundamentally, thesignal processing, especially the signal reconstruction when the A/Dconverter is being overdriven, and the self-test functions are performedanalogously to the above described approach shown in FIG. 2.

As shown in FIG. 4, a pipe segment 10 is surrounded by a first inductivereceiver coil 12 and a second inductive receiver coil 14 which spacedapart from the receiver coil 12 in the axial direction so that a liquid16 flowing in the pipe segment 10 flows through the coils 12 and 14 inthe axial direction. The axial distance of the two coils 12, 14 and theaxial dimensions of the coils 12, 14 are, for example, 2 mm. The tworeceiver coils 12, 14 are surrounded on the outside by a transmittingcoil 18 which is located coaxially to the two coils 12, 14 and has alarger diameter than coils 12, 14. The axial dimensioning of thetransmitter coil 18 is such that the two receiver coils 12, 14 arelocated completely within the transmitter coil 18. Preferably theextension of the transmitter coil 18 in the axial direction is at leasttwice as great as the axial extension of the arrangement of the receivercoils 12, 14, i.e., the distance plus the axial extension of the coils12, 14. The coils 12, 14, 18 are located in a housing 22 which surroundsthe pipe segment 10 and form a probe 11.

Typically, the pipe segment 10 is part of the lubricant circuit of amachine, then the liquid 16, for example, is a lubricant containingmetal particles which are typically abrasion from moving parts of themachine. A typical value for the lubricant flow rate in the main flow is10 liters/min. At much higher flow rates it is advantageous to measure asecondary flow, instead of the main flow rate.

As shown in FIG. 5, the two receiver coils 12, 14 are connectedsubtractively as a difference coil 15, i.e., they are arranged in anopposite direction, so that a voltage with the same amount but withopposite signs is induced in the two coils 12, 14. The transmitter coils18 and the receiver coils 12, 14 form a transformer arrangement, wherethe transmitter coil 18 forms the primary side and the receiver coils12, 14 forms the secondary side. The transformer core is formed by thematerials or media fed through the coils 12, 14, 18, e.g., air, thehousing 22, the pipe 10, and the liquid 16 with the particles 20.

The impedance difference of the coils 12, 14 caused by the particles 20,i.e. the difference of the impedance of the two coils 12, 14 caused bythe instantaneous presence of a particle 20 in one of the two coils 12,14 (the particles 20 are much smaller than the distance of the coils 12,14), is imaged by the measurement signal output by the coils 12 and 14.

FIG. 3 shows one example of the structure of the eddy currentmeasurement device that uses the probe 11 according to an aspect of thepresent invention.

A transmitter coil 18 is used, by means of an alternatingelectromagnetic field with at least one given carrier frequency, toinduce eddy currents in the particles 20, which in turn induce an ACvoltage that acts as the probe signal in the receiving coil 15. Thereceiving coil 15 is a difference coil and with a voltage having acarrier oscillation with the carrier frequency of the transmitter coil18. The amplitude and the phase of the probe signal are modulated by aparticle 20 when the latter travels into the effective width WB of thereceiving coil 15.

The voltage of the transmitter coil 18 can be produced, for example, bya binary signal produced by a timer unit 44 input to a generator 48producing a rectangular signal and a sinusoidal signal, which travelsthrough the curve shaper 40 and then is amplified by a power amplifier42 before being sent to the transmitter coil 18. Preferably the signalhas a sinusoidal shape and in the simplest case contains only a singlecarrier frequency, but maybe several carrier frequencies at the sametime and/or carrier signals which differ distinctly from sinusoidaloscillations. Typically the carrier frequency is in the range from 5 kHzto 1 MHz.

The probe signal received by the receiving coil 15 travels through abandpass filter 19 and an adjustable preamplifier 17 before beingsupplied to an A/D converter stage 35. The bandpass filter 19 is used,on the one hand, by means of a lowpass filter as an (anti-)aliasingfilter with respect to signal digitization by the A/D converter stage35, and on the other hand, by means of a highpass to mask out high andlow frequency noise signals. The adjustable preamplifier 17 is used tobring the amplitude of the analog probe signal to the amplitudeoptimally suitable for the A/D converter stage 35.

The A/D converter stage 35 has two A/D converters 32, 34 which areconnected in parallel and have high resolution with a resolution of 16bits, preferably at least 22 bits, and are able to carry out at least500 A/D conversions per second. The A/D converters 32, 34 are preferablymade as flash converters or SAR (successive approximation register)converters.

If offset voltage compensation takes place by means of an additional D/Aconverter and subtractor, a resolution of the A/D converter of 12 bitsis sufficient.

The A/D converter stage 35 is triggered by a trigger means 37 which hasthe aforementioned timer unit 44, the cosine generator 48, the sinegenerator 46 which is located parallel to the cosine generator 48, andthe frequency divider 30. A signal is provided to the frequency divider30. The signal has been generated by the cosine generator 48 and has thefrequency of the carrier frequency of the supply signal of thetransmitter coil 18, and the signal of the sine generator 46 whichcorresponds to the signal of the cosine generator 48, but which isphase-shifted by 90° to with respect to the signal of the sine generator46. In the frequency divider 30 these two signals are divided withrespect to their frequency by a whole number n. The correspondingfrequency-reduced output signal is used to trigger the A/D converter 32and the A/D converter 34. The selection of the number n for the divider30 is undertaken by a digital signal processor 60 depending on theparticle frequency, i.e., which as the quotient of the flow velocity vof the liquid 16, the velocity v of the particles 20, and the effectivewidth WB of the receiving coil 15. Preferably, n is chosen to beinversely proportional to the particle frequency in order for thetrigger rate of the A/D converter stage 35 to be at least roughlyproportional to the particle frequency. Therefore, if the effectivewidth WB in the first approximation is assumed to be constant, at ahigher flow/particle velocity v and thus higher particle frequency theanalog probe signal is sampled more frequently.

Preferably, the divider 30 is made as a so-called PAL (ProgrammableArray Logic) component in order to ensure that the trigger signalsarrive as synchronously with the output signal of the cosine generator48 and the sine generator 46 and without phase jitter at the A/Dconverter stage 35.

Due to the corresponding phase shift of the two input signals of thedivider 30, the two A/D converters 32, 34 are also triggered with afixed phase offset of 90°. In this way, the analog probe signal can beevaluated in a two-component manner, i.e., both with respect toamplitude and phase. It goes without saying that the phase delay betweenthe trigger signal of the A/D converter signal 35 and the signal of thetransmitter coil 18 should be as small as possible, and especiallyso-called phase jitter should also be avoided, i.e., the phase relationsshould be as constant in time as possible.

It is ensured that the analog probe signal is sampled by each A/Dconverter 32 and 34 at most once per full wave of the carrieroscillation (in this case n is equal to 1) with the illustrated triggermeans 37. Depending on the current fault frequency, i.e., the velocityof the test piece v, n however can be much greater than 1 so thatsampling only takes place at each n-th full wave of the carrieroscillation.

Since sampling takes place at most once per full wave per A/D converter32, 34, the frequency of the carrier oscillation, i.e., the carrierfrequency, is eliminated from the digital signal by this undersampling,i.e., demodulation of the analog probe signal takes place by means ofundersampling.

Preferably, n is chosen such that a noticeable particle signal isobserved in the time interval. That is, a time interval is chosen suchthat one point of a particle 20 moves through the effective width WB ofthe receiving coil 15 the time interval which corresponds essentially tothe inverse of the main particle frequency, which is at least 5,preferably at least 20 samples are taken by each A/D converter 32 and 34to obtain the information contained in the particle signal sufficientfor reliable particle detection. Generally however not more than 50, atmost 100, samplings will be necessary during this time interval, aminimum of 10 samplings.

The frequency of the carrier oscillation should be chosen such that itis at least ten times the particle frequency, since otherwise theparticle signal is carried by too few full waves of the carrieroscillation and the reproducibility of particle detection becomes aproblem.

The demodulated, digital, two-channel output signal of the A/D converterstage 35 travels through a digital bandpass filter 52 may be containedin the signal processor 60 and used to mask out noise signals which areoutside the bandwidth of the particle signal. For this purpose, thecorner frequency of the highpass is preferably chosen such that thecorner frequency is less than one fourth of the particle frequency,while the corner frequency of the lowpass filter is preferably chosensuch that it is at least twice the particle frequency in order to avoidmasking out the signal portions which still contain information withrespect to particle passage.

The digital bandpass filter 52 is clocked with the sampling rate of theA/D converter stage 35, i.e., the trigger rate; this entails the majoradvantage that the corner frequencies of the bandpass filter when theparticle frequency changes, i.e., when the velocity of the particles vchanges, are automatically entrained with the particle frequency sincethe corner frequencies of a digital bandpass filter are proportional tothe clock rate which is automatically adapted to the change of theparticle frequency by way of the sampling rate which is stipulated bythe trigger unit 37.

The information which is necessary for determining the main particlefrequency with respect to the effective width WB can be either inputmanually to the signal processor 60 or provided directly by themeasurement head 11, as is described for example in European PatentApplication No. 0 734 522 B1 and corresponding to International PatentApplication Publication. No. 95/16912.

It goes without saying that the measurement system reacts analogously tothe change of the particle frequency which is caused by the particlevelocity v being kept constant, but the receiving coil 15 is replaced byanother with a different effective width WB.

In particular, for relatively large values of n, i.e., when only arelatively small number of full waves of the carrier oscillation at allis sampled, during the sampling pauses for example the transmitter coil18 and/or the evaluation electronics, i.e., especially the signalprocessor 60, can be turned off or put on stand-by in order to reducepower consumption. This is important especially for portable measurementdevices.

The useful signal obtained after filtration by the digital bandpassfilter 52 is evaluated in an evaluation unit 50 in order to detect thepassage of particles 20 using the amplitude and phase information of theparticle signal.

Advantageously, the evaluation unit 50 is made such that the detectedparticle passages are counted so that conclusions can be made about theparticle concentration in the liquid 16, and the state of the machine.

Fundamentally, in a difference coil, as a result of difference formation(the individual coils of the difference coil are never exactly alike inpractice), the so-called coil offset voltage arises that can exceed theactual fault signal by several orders of amplitude, for example, by 100to 30000 times. The resulting relatively large amplitude of the receivercoil signal compared to the actual useful signal imposes high demands onthe electronics, especially on the A/D converter on its resolution.

Monitoring and calibration functions, which are named above inconjunction with FIG. 1, are implemented in the self-test unit 62 of theprocessor 60. Thus, the self-test unit 62 controls the switcharrangement 66 with three switches 63, 67, 69 in order to feed thesignal for the transmitter coil 18 of the probe 11 by bypassing thetransmitter coil 18 and the receiver coil 15 directly as a periodicinput signal into the signal processing, i.e., into the input of thebandpass filter 19.

What is claimed is:
 1. A device for detecting electrically conductiveparticles in a liquid flowing in a pipe segment with a velocity (v),comprising a transmitter coil arrangement with at least one transmittercoil for exposing the liquid to a plurality of periodic alternatingelectromagnetic fields inducing a plurality of eddy currents in theparticles; a receiver coil arrangement with at least one receiver coilfor detecting a periodic electrical signal, according to the inducededdy currents and which has a carrier oscillation where at least one ofan amplitude and a phase is modulated by the particles when theparticles travel into an effective width of the receiver coilarrangement; a signal processing unit for producing a useful signal froma receiver coil signal; an evaluation unit for evaluating the usefulsignal to detect the passage of electrically conductive particles in apipe segment; and a self-test unit which automatically or upon externalrequest undertakes systematic quantitative checking of at least one of aplurality of signal processing functions of the signal processing unit,the transmitter coil arrangement, and the receiver coil arrangement, andupon external request to undertake calibration of the signal processingunit by means of a calibration standard replaces at least one of thetransmitter coil arrangement and the receiver coil arrangement.
 2. Amethod for detecting electrically conductive particles in a liquidflowing in a pipe segment with a velocity (v), comprising exposing theliquid to periodic alternating electromagnetic fields with a transmittercoil arrangement in order to induce eddy currents in the particles;detecting a periodic electrical signal according to the eddy currentswith a receiver coil arrangement, such that the receiver coil which hasa carrier oscillation whose amplitude and phase are modulated by theparticles when the particles travel into an effective width of thereceiver coil arrangement; producing a useful signal from the receivercoil signal by means of a signal processing unit; evaluating the usefulsignal with an evaluation unit in order to detect a passage ofelectrically conductive particles in the pipe segment; systematicallyand quantitatively checking at least one of the signal processingfunctions of the signal processing unit, the transmitter coilarrangement, and the receiver coil arrangement, wherein the systematicand quantitative checking is undertaken automatically or upon request,and a calibration standard is substituted for at least one of atransmitter coil arrangement and a receiver coil arrangement; andundertaking a calibration of the signal processing unit.